The present invention relates to a method for driving a switch, which controls the current drawn by an inductive energy storage element, in a switched-mode converter which is in the form of a step-up converter, in particular in a switched-mode converter which is used as a power factor correction circuit (Power Factor Controller, PFC), and to a drive circuit for a switch such as this in a switched-mode converter.
A switched-mode converter that is used in a PFC circuit is described, by way of example, in DE 100 40 411 A1. A drive circuit for a switch to control the power consumption in a PFC circuit is the integrated module of the TDA4863 type from Infineon Technologies AG, Munich, which is described in “Boost Controller TDA 4683, Power Factor Controller IC for High Power and Low THD”, Data Sheet, V 1.0, Infineon Technologies AG, May 2003. The use of the integrated module in a power factor correction circuit is described in “TDA—Technical Description AN-PFC-TDA 4863-1”, Application Note, V1.2, Infineon Technologies AG, October 2003.
The basic design of a switched-mode converter such as this will be explained in the following text, with reference to FIG. 1, in order to assist understanding of the problem on which the invention is based.
The object of a switched-mode converter that is used as a PFC is to provide a DC voltage Vn for a load from an AC voltage Vn, in particular a power supply AC voltage, in which case the mean current drawn by the PFC should be at least approximately proportional to the profile of the input voltage Un in order to receive mainly real power.
The switched-mode converter that is illustrated in FIG. 1 has connecting terminals K1, K2 for application of an input voltage Vn, for example a sinusoidal power supply voltage, and a rectified GL which is connected downstream from the input terminals and produces a rectified voltage Vin from the input voltage Vn and the terminals K3, K4. These terminals K3, K4 are also referred to in the following text as input terminals of the switched mode converter. A converter stage with a step-up converter topography is arranged between these input terminals K3, K4 and output terminals KS, K6. In parallel with the input terminals K3, K4, this converter stage has a series circuit comprising an inductive energy storage element L1, for example a storage conductor, and a switch T which, for example, is in the form of a power transistor. A second rectify arrangement, which in the example comprises a diode D and a capacitor C, is connected in parallel with the switch T, and, when the switch T is open, in series with the inductive energy storage element L1. The capacitor C is connected between the output terminals KS, K6, at which an output voltage Vout is available.
In this switched-mode converter, which is in the form of a step-up converter, the inductive energy storage element Ll receives energy when the switch T is closed, and emits this energy to the output capacitor C and to the output terminals KS, K6 when the switch is subsequently opened.
A control signal, which is dependent on the output voltage Vout and is provided by a regulator 10, is available in the switched-mode converter. The regulator 10 forms the difference between this output signal Sout (which is produced by a voltage divider R3, R4 from the output voltage Vout) and a reference value Vref, and produces the control signal S10 as a function of this difference. In the simplest case, the regulator comprises an operational amplifier 11, which is also referred to as an error amplifier and is connected externally to an impedance Z in order to adjust the control response.
In order to produce a drive signal S20 for the switch T, the control signal S10 is multiplied by an input signal Sin, which is dependent on the rectified input voltage Vin and is produced by means of a voltage divider R1, R2, C1 from the input voltage Vin, in order to produce a comparison signal S21 which is supplied to a drive signal production circuit 20.
This signal production circuit 20 produces a pulse-width-modulated drive signal S20 in order to drive the switch T and is designed to always produce a switching-on level for the drive signal S20, in order to switch on the switch T, as soon as the storage inductor is free of energy after the switch has been switched off, that is to say when the drive signal S20 is at a switching-off level. An auxiliary winding is used to determine the storage states in which the storage inductor is free of energy, and is inductively coupled to the inductor L1 and supplies a magnetization signal S22 to the signal production circuit 20, with this magnetization signal S22 indicating the magnetization state of the storage inductor L1.
In order to adjust the switched-on duration, the signal production circuit 20 compares the comparison signal S21 (which depends on the input voltage Vin and the control signal S10) with a current measurement signal S23 which is dependent on the current through the switch T. The current through the switch T, and thus the current measurement signal S23 rise, when the switch T is closed, in proportion to the input voltage Vin. A switching-off level for the drive signal S20 is produced by the signal production circuit 20 once the current measurement signal S23 has risen to the value of the comparison signal.
FIG. 2a illustrates the waveform of the current measurement signal S23 for two successive drive cycles.
The profile of the input current Iin is also shown, by dashed lines, with this input current Iin corresponding to the current through the switch T during the period in which the drive signal S20 is switched on and falling to zero during the period in which it is switched off, which is equivalent to demagnetization of the inductor L1. For the illustration in FIG. 3, the value of the measurement signal S23 corresponds to the input Iin, whose peak value is limited by the comparison signal S21.
FIG. 2b shows the profile of the drive signal S20, which is formed as a function of the magnetization state of the inductor and as a function of a comparison between the current measurement signal S23 and a comparison signal. Ton in this case denotes the switched-on duration, during which the drive signal S20 assumes a switching-on level for the switch T, and Toff denotes a switched-off duration, during which the drive signal S20 assumes a switching-off level.
FIG. 3 shows the waveform of the input voltage Vin for one period of a rectified input voltage Vin in the form of the magnitude of a sine wave, the profile which results from this of the comparison signal S21 in the presence of a control signal S10 that is assumed to be constant for this period, and the profile of the input current. The relationship between the comparison signal S21 and the input voltage Vin results in the comparison signal S21 likewise rising when the input voltage Vin rises. Since the current through the switch T likewise rises as the input voltage Vin rises, constant switched-on durations Ton ideally result when the control signal S10 remains the same, that is to say when the load conditions at the output remain the same, while the switched-off durations Toff vary. The mean value of the input current Iin is in this case proportional to the input voltage.
It can be shown that, for the instantaneous value of the power consumption of a power factor correction circuit such as this:P=0.5·Vin2·Ton/L1  (1a)
Furthermore, the power consumption can also be indicated using the relative switched-on duration d=Ton/T:P=0.5·Vin2·d·T/L1=0.5·Vin2·d/(L1·f)  (1b).
In this case, P denotes the instantaneous value of the power consumption, Vin the input voltage, Ton the switched-on duration, L1 the inductance value of the inductor, and f=1/T the switching frequency. The above relationships for the instantaneous value of the power consumption P are also valid when the overall period duration T is not constant.
From (1a), the switched-on duration Ton is obtained as a function of the input current Iin as follows:Ton=Îin·L/Vin  (2a),in a corresponding manner, the relative switched-on duration d is:d=Îin·L/(Vin·T)  (2b)where Îin denotes the peak value of the input current Iin reached in each drive cycle. This peak value is proportional to the comparison signal S21, so that:Ton=k·S21·L/Vin  (3)where k denotes a proportionality factor. Substitution of (3) in (1) gives:P=0.5·k·S21·Vin  (4)
As is evident from (1), the switched-on duration Ton for a given power consumption is inversely proportional to the square of the input voltage Vin. For so-called wide-range power supply units which have to be designed to produce a constant output voltage Vout for input voltages Vin with peak values between 90V and 270V, this means that the switched-on duration for an input voltage of 90V (=⅓·270V) must be 9 times the switched-on duration for a voltage of 270V. The comparison signal S21 for a given power consumption is inversely proportional to the input voltage Vin. During one period of the input voltage, the power consumption is in each case a maximum when the input voltage Vin reaches its maximum value. The comparison signal S21 is also maximized at this time. If one considers the range over which peak values of the input voltage Vin can fluctuate, then the comparison signal assumes its maximum value at the peak value of the smallest possible input voltage.
In the case of a wide-range power supply unit, it is assumed that the rated power consumption is reached when the input voltage Vin assumes a peak value 90V and the comparison signal S21 assumes a maximum value S21max. If the input voltage changes such that peak values of 270V occur, then the maximum value of the comparison signal S21 is reduced to S21max/3. This comparison signal S21 is matched to different input voltage conditions by means of the control signal S10.
If an overload now occurs at the output of the converter when the input voltage is high, resulting in the output voltage Vout falling, then the comparison signal S21 can be regulated up to its maximum value S21max by means of the regulator 10, thus resulting in a power consumption which corresponds to 9 times the rated power. This can lead to instabilities in the power consumption control process.
One aim of the present invention is to provide a method for driving a switch which controls the power consumption in a power factor correction circuit, which method ensures stable control of the power consumption, and to provide a drive circuit for driving a switch in a power factor correction circuit.